Doherty amplifier circuits

ABSTRACT

A Doherty amplifier circuit comprising: a splitter having: a splitter-input-terminal for receiving an input signal; a main-splitter-output-terminal; and a peaking-splitter-output-terminal; a main-power-amplifier having a main-power-input-terminal and a main-power-output-terminal, wherein; the main-power-input-terminal is connected to the main-splitter-output-terminal; and the main-power-output-terminal is configured to provide a main-power-amplifier-output-signal; a peaking-power-amplifier having a peaking-power-input-terminal and a peaking-power-output-terminal, wherein: the peaking-power-input-terminal is connected to the peaking-splitter-output-terminal; and the peaking-power-output-terminal is configured to provide a peaking-power-amplifier-output-signal. The splitter, the main-power-amplifier and the peaking-power-amplifier are provided by means of an integrated circuit.

The present disclosure relates to Doherty amplifier circuits, and inparticular to Doherty amplifier circuits that can, at least in part, beimplemented as an integrated circuit.

According to a first aspect of the present disclosure there is providedA Doherty amplifier circuit comprising:

-   -   a splitter having:        -   a splitter-input-terminal for receiving an input signal;        -   a main-splitter-output-terminal; and        -   a peaking-splitter-output-terminal;    -   a main-power-amplifier having a main-power-input-terminal and a        main-power-output-terminal, wherein:        -   the main-power-input-terminal is connected to the            main-splitter-output-terminal; and        -   the main-power-output-terminal is configured to provide a            main-power-amplifier-output-signal;    -   a peaking-power-amplifier having a peaking-power-input-terminal        and a peaking-power-output-terminal, wherein:        -   the peaking-power-input-terminal is connected to the            peaking-splitter-output-terminal; and        -   the peaking-power-output-terminal is configured to provide a            peaking-power-amplifier-output-signal;    -   an integrated circuit;    -   wherein the splitter, the main-power-amplifier and the        peaking-power-amplifier are provided by means of the integrated        circuit.

In one or more embodiments the splitter-input-terminal is connected tothe main-splitter-output-terminal. The splitter may comprises asplitter-attenuator and a splitter-phase-shifter connected in cascadebetween the splitter-input-terminal and thepeaking-splitter-output-terminal.

In one or more embodiments the splitter-attenuator and thesplitter-phase-shifter are provided on the integrated circuit.

In one or more embodiments the splitter-attenuator comprises a variablesplitter-attenuator that is configurable to apply a variable attenuationor amplification factor to the input signal.

In one or more embodiments the variable splitter-attenuator comprises: afirst-variable-resistor; and a second-variable-resistor. Thesecond-variable-resistor may be connected between anattenuator-input-terminal and an attenuator-output-terminal. Thefirst-variable-resistor may be connected between theattenuator-input-terminal and a reference-terminal.

In one or more embodiments the splitter-phase-shifter comprises avariable splitter-phase-shifter that is configurable to apply a variablephase shift to the input signal.

In one or more embodiments the variable splitter-phase-shiftercomprises: a first-variable-capacitor, a first-inductor, asecond-variable-capacitor, a third-variable-capacitor, asecond-inductor, and a fourth-variable-capacitor. Thefirst-variable-capacitor may be connected between aphase-shifter-input-terminal and a reference-terminal. Thefirst-inductor may be connected between the phase-shifter-input-terminaland an intermediate-node. The second-variable-capacitor may be connectedbetween the intermediate-node and the reference-terminal. Thethird-variable-capacitor may be connected between the intermediate-nodeand the reference-terminal. The second-inductor may be connected betweenthe intermediate-node and a phase-shifter-output-terminal. Thefourth-variable-capacitor may be connected between thephase-shifter-output-terminal and the reference-terminal.

In one or more embodiments the Doherty amplifier further comprises acontroller that is configured to:

-   -   receive one or more sensed-temperature-signals; and    -   set control parameters of the variable splitter-attenuator        and/or the variable splitter-phase-shifter based on the        sensed-temperature-signals.

In one or more embodiments the sensed-temperature-signals arerepresentative of: a temperature of the integrated circuit, or atemperature of the main-power-amplifier or the peaking-power-amplifier.

In one or more embodiments the Doherty amplifier further comprises: atransformer having a transformer-input-terminal and atransformer-output-terminal. The transformer-input-terminal may beconnected to the peaking-power-output-terminal. Thetransformer-output-terminal may be configured to provide atransformer-output-signal to a combining node.

In one or more embodiments the transformer is provided on the integratedcircuit.

In one or more embodiments the Doherty amplifier circuit furthercomprises:

-   -   a package; and    -   a laminate;    -   wherein the integrated circuit and the laminate are provided in        the package.

In one or more embodiments the Doherty amplifier circuit furthercomprises a main-output-impedance-inverter connected between themain-power-output-terminal and a combining node. Themain-output-impedance-inverter may comprise a CLC-inductor, afirst-CLC-capacitor and a second-CLC-capacitor. The CLC-inductor may beconnected between the main-power-output-terminal and the combining node.The first-CLC-capacitor may be connected between themain-power-output-terminal and a reference terminal. Thesecond-CLC-capacitor may be connected between the combining node and thereference terminal.

In one or more embodiments the first-CLC-capacitor may be provided onthe integrated circuit. The CLC-inductor and the second-CLC-capacitormay be provided on the laminate.

In one or more embodiments the integrated circuit comprises a BiCMOScircuit.

While the disclosure is amenable to various modifications andalternative forms, specifics thereof have been shown by way of examplein the drawings and will be described in detail. It should beunderstood, however, that other embodiments, beyond the particularembodiments described, are possible as well. All modifications,equivalents, and alternative embodiments falling within the spirit andscope of the appended claims are covered as well.

The above discussion is not intended to represent every exampleembodiment or every implementation within the scope of the current orfuture Claim sets. The figures and Detailed Description that follow alsoexemplify various example embodiments. Various example embodiments maybe more completely understood in consideration of the following DetailedDescription in connection with the accompanying Drawings.

One or more embodiments will now be described by way of example onlywith reference to the accompanying drawings in which:

FIG. 1 shows an example embodiment of a Doherty amplifier circuit;

FIG. 2 shows another example embodiment of a Doherty amplifier circuit;

FIG. 3 shows an example implementation of a main-power-amplifier or apeaking-power-amplifier;

FIG. 4 shows a simplified schematic of a driver that can be used in FIG.3;

FIG. 5 shows a capacitively loaded gyrator that can provide low-ohmicbiasing for a driver;

FIG. 6 shows schematically an analysis of an inter-stage-matchingnetwork between a pre-driver and a driver;

FIG. 7 shows a simplified schematic of a pre-driver;

FIG. 8 shows an example configuration of a splitter circuit;

FIGS. 9a to 9c show various implementations of circuits that can be usedto provide the functionality of a variable component that is disclosedherein; and

FIG. 10 shows an alternative implementation of a splitter-phase-shifter.

FIG. 1 shows an example embodiment of a Doherty amplifier circuit 100.The Doherty amplifier 100 has a Doherty-input-terminal 102 and aDoherty-output-terminal 104. Amongst other things, a splitter 106, amain-power-amplifier 112 and a peaking-power-amplifier 114 are connectedbetween the Doherty-input-terminal 102 and the Doherty-output-terminal104. As will be discussed in detail below, advantageously, each of thesecomponents 106, 112, 114 is provided by means of an integrated circuit126.

Some of the components of the Doherty amplifier circuit 100 areimplemented on an integrated circuit (IC) 126, and some of thecomponents are implemented on a laminate. This can result in a compactsolution. The laminate and the IC 126 can together be considered as partof the same Doherty amplifier device. Therefore, the components of theDoherty amplifier circuit 100 can be partitioned between the IC 126 anda laminate-based PA module (such as a dual-layer laminate). Partitioningthe design between the chip and the laminate can be a cost effectiveimplementation, for example in terms of package size and surface mounteddevices (SMD) component count.

The Doherty amplifier circuit 100 has a main path and a peaking path,connected in parallel with each other between the Doherty-input-terminal102 and a combining node 122. The combining node 122 is connected to theDoherty-output-terminal 104 through a matching network 124.

The splitter 106 has a splitter-input-terminal 128, amain-splitter-output-terminal 130 and a peaking-splitter-output-terminal132. The splitter-input-terminal 126 is connected to theDoherty-input-terminal 102 in order to receive an input signal. Thesplitter-input-terminal 126 is also connected directly to themain-splitter-output-terminal 130. In this way, the input signalreceived at the Doherty-input-terminal is provided directly to themain-splitter-output-terminal 130, without any significanttransformation or processing.

The splitter 106 includes a splitter-attenuator 108 and a splitter-phaseshifter 110 connected in cascade between the splitter-input-terminal 128and the peaking-splitter-output-terminal 132. The splitter-attenuator108 and the splitter-phase shifter 110 in this example are variablecomponents, which can be set such that together they apply anattenuation and phase shift to the received input signal before it isprovided to the peaking-splitter-output-terminal 132. It will beappreciated that the splitter-attenuator 108 and the splitter-phaseshifter 110 may be connected between the splitter-input-terminal 128 andthe peaking-splitter-output-terminal 132 in the reverse order to thatshown in FIG. 1, whilst still providing the required functionality.

The main-power-amplifier 112 has a main-power-input-terminal and amain-power-output-terminal. The main-power-input-terminal is connectedto the main-splitter-output-terminal 130. The main-power-output-terminalprovides a main-power-amplifier-output-signal, which in this example isan amplified version of the input signal received at theDoherty-input-terminal 102. Only signals that are larger than a certainthreshold level may be amplified by the peaking-power-amplifier 114.

The peaking-power-amplifier 114 has a peaking-power-input-terminal and apeaking-power-output-terminal. The peaking-power-input-terminal isconnected to the peaking-splitter-output-terminal 132. Thepeaking-power-output-terminal provides apeaking-power-amplifier-output-signal, which in this example is anamplified version of the phase-shifted and attenuated input signalreceived at the Doherty-input-terminal 102.

As indicated above, the splitter 106, the main-power-amplifier 112 andthe peaking-power-amplifier 114 are provided on the IC 126. This can beconsidered as monolithically integrating the components. Thisbeneficially enable a physically small and compact circuit to beprovided. Also advantageously, the properties of the splitter 106 can betailored to the downstream components of the Doherty amplifier 100. Forexample, the splitter 106 does not have to be implemented for a worstcase scenario of the downstream components such as the main- andpeaking-power-amplifiers 112, 114 (power trains) and a Doherty combiner120. Therefore, the Doherty amplifier circuit 100 as a whole can beconsidered more efficient and smaller. Further still, by integrating thesplitter 106 and power-amplifiers 112, 114 on the integrated circuit126, a fixed relative position between these components will be known tothe circuit designer. Therefore, their expected interaction with eachother can be more accurately modelled and the Doherty amplifier circuit100 can be particularly efficient. This may not be achievable if thesplitter 106 and power-amplifiers 112, 114 were not integrated becausetheir relative positions would not be definitively known, nor would theynecessarily be fixed. A yet further advantage is that the splitter 106and the power-amplifiers 112, 114 can both make use of internal circuitsthat are available on the IC 126, such as power supply circuits.Therefore, the need for an external power regulator to provide a stablevoltage supply for one or more of the splitter 106 and thepower-amplifiers 112, 114 can be reduced or avoided.

Also, as will be discussed in more detail below, by implementing thesplitter 106 and the power-amplifiers 112, 114 on the integrated circuit126, better temperature control can be achieved, and better calibration,even self-calibration, of the Doherty amplifier circuit 100 can beachieved. A further advantage of monolithic integration is a goodmatching of the splitter 106 and the power amplifiers 112 and 114.Furthermore, splitter integration can provide the possibility forreduction of the spread on main parameters over process by the choice ofoptimal input amplitude and phase settings via a SPI (serial peripheralinterface) or an OTP (one-time programmable) memory cell.

In some applications, the integrated circuit can be implemented withBiCMOS technology. In some other technologies, it may not be possible toimplement the components of the splitter 106 and the power-amplifiers112, 114 on the same IC 126.

In this example, the Doherty amplifier circuit 100 includes atransformer 118 in the peaking path. The transformer 118 may also bereferred to as a peaking transformer. The transformer 118 is used toperform impedance transformation, and can advantageously enable widebandamplifier operation. This can be achieved because the transformer 118can have low losses and/or a low leakage inductance.

The transformer 118 has a transformer-input-terminal and atransformer-output-terminal. The transformer-input-terminal is connectedto the peaking-power-output-terminal of the peaking-power-amplifier 114.The transformer-output-terminal provides a transformer-output-signal tothe combining node 122. The transformer 118 in this example comprisestwo magnetically coupled inductors, and can be implemented as: (i) anautotransformer with two sub-windings; or (ii) a standard (real)transformer with a primary winding and a secondary winding.

In examples where an autotransformer is used, it may comprise afirst-end-terminal, a second-end-terminal and an intermediate-terminal.The first-end-terminal may be galvanically connected to the combiningnode 122. The intermediate-terminal may be galvanically connected to thepeaking-power-output-terminal. The second-end-terminal may be connectedto a reference terminal (not shown).

In examples where a standard (real) transformer is used, a first-windingmay be galvanically connected between the peaking-power-output-terminaland a reference terminal. A second-winding may be galvanically connectedbetween the combining node 122 and the reference terminal.

In this implementation, the transformer 118 is also provided on theintegrated circuit 126. In other implementations, the transformer 118can be provided on the laminate (not shown).

A main-output-impedance-inverter 116 is connected between themain-power-output-terminal of the main-power-amplifier 112 and thecombining node 122. The main-output-impedance-inverter 116 provides animpedance inversion and a 90 degrees phase shift.

In this example, the main-output-impedance-inverter 116 is provided onthe laminate and not on the IC 126. However, as will be discussed below,one or more sub-components of the main-output-impedance-inverter 116 canbe provided on the IC 126 in other examples.

The splitter-attenuator 108 and the splitter-phase shifter 110, which inthis case are placed in front of the peaking-power-amplifier 114, areused such that the signals that arrive at the combining node 122 fromthe main path and the peaking path are in phase with each other.Therefore, in FIG. 1, there is no need to compensate for an additional90 degrees in the peaking path, downstream of thepeaking-power-amplifier, as may be the case for other Doherty amplifiercircuits.

In other cases, the splitter-attenuator 108 and the splitter-phaseshifter 110 may be placed in front of the main-power-amplifier 112; orthe splitter-attenuator 108 and the splitter-phase shifter 110 can beplaced in front of both the main- and peaking-power-amplifier 112, 114.

In some examples, the splitter-attenuator 108 can apply anattenuation-factor or an amplification factor that is between 0 and 1such that it amplifies a received signal. In such examples, thesplitter-attenuator 108 can be referred to as a splitter-amplifier.

The main-output-impedance-inverter 116 and the transformer 118 can beconsidered together as a wide-band Doherty combiner 120.

FIG. 2 shows another example embodiment of a Doherty amplifier circuit200, which is shown as a block diagram of a possible Doherty-amplifierline-up. Components that are also illustrated in FIG. 1 have been givencorresponding reference numbers in the 200 series, and will notnecessarily be described again here.

The impedance levels shown are the real parallel-equivalent parts ofinput- or load impedances (impedances seen when looking to the right).

Both the main branch (at the top) and the peaking branch (at the bottom)contain three amplifier stages. That is, the main-power-amplifier 212includes, from left to right: a main-pre-driver 212 a, a main-driver 212b, and a main-final-stage (or main-output-stage) 212 c. Similarly, thepeaking-power-amplifier 214 includes: a peaking-pre-driver 214 a, apeaking-driver 214 b, and a peaking-final-stage (orpeaking-output-stage) 214 c.

An input splitter 206 is shown, with which the amount of attenuation andphase shift in the peaking branch can be controlled by the user in orderto improve or optimize the overall linearity of the Doherty amplifier.The default phase shift introduced by the splitter-phase shifter 210(which is in front of the peaking-power-amplifier chain 214) is around90 degrees, which cancels with the 90 degrees phase shift introduced bythe main-output-impedance-inverter 216 following themain-power-amplifier chain 212. In this way, constructive signaladdition is achieved at the combining node 222.

The power of the input signal received at the Doherty-input-terminal(input power) is split into two equal parts in this example because boththe main- and the peaking-power-amplifier chains 212, 214 have a 100Ωinput resistance. Therefore the overall input resistance is(100//100)=50Ω. At the right-hand side we encounter the output combiner220, followed by a common output-matching network 224. In this example,the common output-matching network 224 is meant for a 50-Ω load.

The main-output-impedance-inverter 216 in this example is implemented asa CLC circuit (not shown). The CLC circuit includes a CLC-inductor, afirst-CLC-capacitor and a second-CLC-capacitor. The CLC-inductor may beconnected between the main-power-output-terminal of themain-power-amplifier 212 and the combining node 222. Thefirst-CLC-capacitor may be connected between themain-power-output-terminal and a reference terminal. Thesecond-CLC-capacitor may be connected between the combining node 222 andthe reference terminal. The first-CLC-capacitor in this example isprovided on the IC. The CLC-inductor and the second-CLC-capacitor areprovided on the laminate in this example. In this way, the main branchof the output combiner 220 contains a mostly off-chip on-laminate lumpedCLC impedance inverter (as opposed to a quarter-wavelength transmissionline), in order to save area.

As discussed above, the peaking branch of the output combiner 220contains an on-chip output transformer 218. In some examples, the entireoutput combiner 220 can be implemented on the laminate.

FIG. 3 shows an example implementation of a power-amplifier 312, whichcan be used as the main-power-amplifier or the peaking-power-amplifierof FIG. 2. For example, the main- and peaking-power-amplifier chains canhave the same 3-stage topology as shown in FIG. 3, but they may bedimensioned differently. The impedance levels within themain-power-amplifier chain in one example are roughly a factor[(1−a)/a]=5/3 higher with respect to those within the peaking amplifierchain. Here parameter a equals ⅜ which corresponds to a back-off levelof 8.5 dB, which can be suitable for amplification of signals having apeak-to-average-power ratio (PAPR) of around 8 dB.

The input resistance of the amplifier chain 312 is approximately equalto the resistance of resistor R1 334, since the input stage of thepre-driver 312 a acts as virtual ground in good approximation due toparallel feedback via resistor R2 336. The resistor ratio −R2/R1 definesthe voltage gain of the pre-driver 312 a, assuming high loop-gain.

The inter-stage matching network between the pre-driver 312 a and thedriver 312 b consists out of an autotransformer L1 338 and a seriesmatching capacitor C1 340. The autotransformer L1 338 can halve theoutput voltage swing of the pre-driver 312 a, but can double its currentdrive capability. In other words, it can perform a factor of 4 impedancedown transformation. The series matching capacitor C1 340 can convertthe signal voltage into a signal current (trans-admittance jωC1) whichdrives the driver stage 312 b. In this way it can perform currentsteering, which is beneficial for the linearity of the driver 312 b.

The driver 312 b converts this input current into an output voltageaccording to trans-impedance −1/(jωC2). Hence the voltage gain aroundthe driver 312 b equals −C1/C2 in good approximation. Since the driver312 b should clip before the pre-driver 312 a clips in case ofincreasing signal amplitude, (C1/C2) can be set such that it is greaterthan 2. Parasitic collector-base capacitance of the driver 312 b is thedominant contribution to C2 342. Capacitive loading on the output ofpre-driver stage 312 a S1, due to C1 340 and parasitics, can be tunedout by the magnetization inductance of autotransformer L1 338. Adetailed description of the inter-stage matching between the pre-driver312 a and the driver 312 b will be provided below under the heading“Driver: Inter-stage matching between pre-driver and driver”.

The inter-stage matching network between the driver 312 b and the finalstage 312 c consists out of an autotransformer L2 344, a DC-blockingcapacitor C3 346, a matching coil L3 348, and a series matchingcapacitor C4 350. The autotransformer L2 344 can halve the outputvoltage swing of the driver 312 b but can double its current drivecapability. In other words, it can perform a factor of 4 impedance downtransformation. The DC-blocking capacitor C3 346 can serve as a DC blockand at the same time it can tune out the leakage inductance of theautotransformer L2 344. The series matching capacitor C4 350 can convertthe signal voltage into a signal current (trans-admittance jωC4), whichdrives the final stage 312 c. In this way it can perform currentsteering, which is beneficial for the linearity of the final stage 312c. The final stage 312 c converts this input current into an outputvoltage according to trans-impedance −1/(jωC5). Hence, the voltage gainaround the final stage 312 c equals −C4/C5 in good approximation. Sincethe final stage 312 c should clip before the driver 312 b clips in caseof increasing signal amplitude, (C4/C5) can be set such that it isgreater than 2. Parasitic collector-base capacitance of the final stage312 c is the dominant contribution to C5 352. Capacitive loading on theoutput of the driver stage 312 b S2, due to the series matchingcapacitor C4 350, can be tuned out by the inductance of matching coil L3348. Additional capacitive loading on the output of driver stage 312 bS2, due to its own output capacitance and other parasitics, can be tunedout by the magnetization inductance of autotransformer L2 344.

According to the above discussion, the total voltage gain of the 3-stageamplifier chain 312 can be written as:

$\begin{matrix}{G_{v} = {{- \frac{1}{4}} \cdot \frac{R_{2}}{R_{1}} \cdot \frac{C_{1}}{C_{2}} \cdot \frac{C_{4}}{C_{5}}}} & (1)\end{matrix}$

according to an assumption of high loop gain. Therefore the total powergain can be written as:

$\begin{matrix}{G_{p} = {\frac{R_{1}}{R_{L}}G_{v}^{2}}} & (2)\end{matrix}$

where R_(L) represents the parallel-equivalent resistive part of theload impedance.

PA Driver Design

FIG. 4 shows a simplified schematic of a driver 412 b that can be usedin FIG. 3.

The voltage gain of the driver 412 b is mainly determined by thecapacitance ratio −Cm/Cbc. The circuit is biased via an electronicinductor 454, as will be discussed below under the heading “Driver:Biasing”. Advantageously, the use of electronic inductors can reducecost and assist in miniaturisation, when compared with passiveinductors.

Driver: Dimensioning of the Signal Transistors

Simulations show that the power gain of an output stage, taking intoaccount the losses of an inter-stage-matching network, amounts toapproximately 13 dB at full power. If we want to have the driver around4 dB backed-off with respect to the output stage, such that the overalldistortion will be dominated by the output stage and not by the driver,then the driver should be scaled down with respect to the output stageby not more than (13−4)=9 dB. This corresponds to a scaling-down factorof not more than 10^((9/10))=8.

Driver: Biasing

FIG. 5 shows a capacitively loaded gyrator 556 that can providelow-ohmic biasing for a driver. The gyrator is implemented by means of atransistor 558 and a resistor R 560 between its base and collector. Acapacitive load C 562 is between the base and emitter of thegyrator-transistor 558 (with a transconductance represented by g_(m)).This capacitively loaded gyrator presents a one-port, which behaves as aseries connection of a voltage source (V_(be)), an inductanceL_(eq)=RC/g_(m), and a small resistance 1/g_(m). The equivalentinductance value L_(eq) is chosen such that it has a high impedance forRF and a low impedance for the modulation frequencies to avoid memoryeffects. The small resistance 1/g_(m) can prevent or reduce problemswhich could be caused by avalanche of the amplifier-transistor.Fortunately this resistance 1/g_(m) gets smaller for larger avalanchecurrents and therefore can provide good protection.

Bias current control can be implemented individually for all amplifier,driver and pre-driver stages, in both the main and peaking path of aDoherty amplifier.

Driver: Inter-Stage Matching Between Pre-Driver and Driver

FIG. 6 shows schematically an analysis of an inter-stage-matchingnetwork between a pre-driver and a driver.

The inter-stage-matching network consists of an autotransformer 638(coil with tap, such as a centre tap) in the pre-driver's collectorlead, and a series matching capacitor Cm 640, as shown on the left-handside of FIG. 6.

Although the autotransformer tap does not necessarily need to be in thecentre, in the following piece of text it has been assumed that the tapis exactly in the centre.

For the purpose of analysis, the autotransformer on the left-hand sideof FIG. 6 is replaced by an ideal transformer (2:1) together withmagnetization inductance L_(m) and leakage inductance L_(s), as shown onthe right-hand side. Also, the matching capacitor C_(m) on the left-handside is replaced by a series connection of capacitances C_(s) andC′_(m), as shown on the right-hand side.

The impedance of the leakage inductance L_(s) at the output (centre tap)of the autotransformer cancels with a part (corresponding with C_(s)) ofthe impedance of C_(m). Let's represent the corresponding capacitance ofthe remaining part of the impedance of C_(m) with C′_(m). This C′_(m)together with the collector-base capacitance C_(bc) of the driverdefines the voltage gain of the driver: Av=−C′_(m)/C_(bc) in case oflarge loop gain. Since the autotransformer introduces a voltage divisionof a factor of 2, the magnitude of the driver's voltage gain |A_(v)|should be larger than 2 if the driver's collector voltage should clipbefore the pre-driver's collector voltage clips, in case of increasingsignal amplitude. However, an unnecessary large voltage gain is notdesired since this decreases the pre-driver's efficiency and increasesthe amplitude of the signal current that should be delivered by thepre-driver. Therefore a voltage-gain magnitude |A_(v)| of about 3 is agood choice for some applications. This consideration determines thevalue of matching capacitor C_(m). The capacitance seen on the collectorof the pre-driver is the series connection of C′_(m) and the inputcapacitance C_(in) of the driver divided by 4 due to the impedancetransformation of the autotransformer: (C′_(m)//C_(in))/4 where thesymbol // represents series connection of capacitances. In parallel withthat capacitance (C′_(m)//C_(in))/4 is the output capacitance C_(out) ofthe pre-driver. The sum of these two capacitances,C_(out)+[(C′_(m)//C_(in))/4], is to be tuned out by the magnetisationinductance L_(m) of the autotransformer.

PA Pre-Driver Design

FIG. 7 shows a simplified schematic of a pre-driver 712 a.

For the output stage and driver, the voltage gain is mainly determinedby a capacitance ratio: A_(v)≈−C′_(m)/C_(bc) where C′_(m) represents theeffective capacitance of the series matching capacitor C_(m) taking intoaccount the effect of the series leakage inductance. However, in case ofthe pre-driver, as shown in FIG. 7, the voltage gain is mainlydetermined by a resistance ratio: A_(v)≈R_(pfb)/R_(in,s) where R_(pfb)765 represents the parallel-feedback resistor and R_(in,s) 767represents the resistor in series with the pre-driver's input, whichalso serves as input matching.

Pre-Driver: Input Resistance of Main- and Peaking Pre-Driver

Both the main- and peaking-pre-driver have an input resistance of 100Ωin this example. The programmable attenuator, which is in front of thepeaking chain as shown in FIGS. 1 and 2, is designed such that its inputresistance is 100Ω irrespective of its programmed setting. However itexpects a load resistance of 100Ω. The characteristic impedance of thelumped-element phase shifter (also shown in FIGS. 1 and 2) is 100Ω so aload resistance of 100Ω (input resistance of peaking pre-driver)translates into an input resistance of 100Ω (load resistance forattenuator). Since the inputs of the main- and peaking-branch areconnected in parallel, an overall input resistance of (100//100)=50Ω isobtained. In order to accurately obtain a pre-driver's input resistanceof 100 Ω, 80% of this input resistance is implemented in a passive wayand only 20% in an active way in this implementation. This means aseries resistor of 80Ω together with a non-perfect virtual ground of20Ω. Assuming a current division around the output of a factor of 2(only 50% of the collector signal current is fed back and the rest goesto the next stage), a transconductance g_(m) of 2/(20Ω)=0.1 A/V isrequired. At 60° C., V_(T) equals 29 mV, and so this requires acollector bias current of g_(m)*V_(T)=3 mA.

Pre-Driver: Biasing

Like the driver, the pre-driver is low-ohmicly biased by a capacitivelyloaded gyrator. Details of such a gyrator are provided above under theheading “Driver: Biasing”.

Adjustable Input Power Splitter

FIG. 8 shows an example configuration of a splitter circuit 826.

As discussed above with reference to FIGS. 1 and 2, the splitter 826,which may also be referred to as an input power splitter, splits aninput signal for the main and peaking path. Because the signal in themain path travels through an impedance inverter with 90 degrees phaseshift, the signal in the peaking path also needs 90 degrees phase shift.This phase shift in the peaking path is implemented in the input powersplitter. Both phase and amplitude in the peaking path of the splittercan be adjusted for optimal performance of the Doherty PA. Thisprogrammability of the amplitude controller (implemented as anattenuator 808) and phase controller 810 in the peaking path can be usedto operate under different but optimal settings for differentfrequencies, different temperatures, different supply voltages and evendifferent products as a result of process spread.

In FIG. 8, a splitter-attenuator 808 and a splitter-phase-shifter 810are connected in cascade with each other between aDoherty-input-terminal 802 and a peaking-power-input-terminal of apeaking-power-amplifier 814. The splitter-attenuator 808 is configurableto apply a variable attenuation factor or amplification factor to theinput signal before it is processed by the peaking-power-amplifier 814.The splitter-phase-shifter 810 is configurable to apply a variable phaseshift to the input signal before it is processed by thepeaking-power-amplifier 814. It will be appreciated that the order ofthese components 808 and 810 might be reversed.

In particular, the values of the variable splitter-attenuator 808 and/orthe variable splitter-phase-shifter 810 can be set to reduce thelikelihood of the operation of the Doherty amplifier becoming unstablewhen the temperature of the circuit changes. For example, a controller(not shown) can receive one or more sensed-temperature-signalsrepresentative of a temperature of part of the circuit. For example,sensed-temperature-signals can be received that are representative of: atemperature of the IC, or a temperature of the main- orpeaking-power-amplifier of the Doherty amplifier. The controller canthen set control parameters of one or more variable components withinthe variable splitter-attenuator 808 and/or the variablesplitter-phase-shifter 810 based on the sensed-temperature-signals. Forexample, if the control is digitally implemented, then if thesensed-temperature-signals passes a certain temperature threshold, thecontroller can cause an extra unit capacitor to be added to a CLCimplementation of the splitter-phase-shifter 810 and/or can cause a unitshunt resistor to be removed from the variable splitter-attenuator 808.Such adding and removing can be implemented by means of MOSFET switches,for example.

In one example, the controller can receive sensed-temperature-signalsover a serial interface. The controller can also receive anoperational-signal that is used to set whether or not the controllerapplies an on-chip calibration loop. If the on-chip calibration loop isto be applied, then the controller can use a look-up-table (LUT) todetermine appropriate control parameters for the variablesplitter-attenuator 808 and/or the variable splitter-phase-shifter 810based on the received sensed-temperature-signals. The information in theLUT can be hard-coded, or can be programmable and stored in OTP(one-time programmable) memory, for example. The use of OTP memory canbe beneficial because, for each product, its performance can be measuredin a factory, and then control parameters appropriate for the measuredperformance can be stored in the OTP memory for future use.

In another example, the controller can apply an algorithm to thereceived sensed-temperature-signals in order to determine controlparameters that are to be applied to the variable splitter-attenuator808 and/or the variable splitter-phase-shifter 810.

The splitter-attenuator 808 includes a first-variable-resistor 866 and asecond-variable-resistor 868. The second-variable-resistor 868 isconnected between an attenuator-input-terminal and anattenuator-output-terminal. The first-variable-resistor 866 is connectedbetween the attenuator-input-terminal and a reference-terminal 876. Thevalues of the first-variable-resistor 866 and/or thesecond-variable-resistor 868 can be set/adjusted in order to provide adesired attenuation factor, and therefore control the magnitude of thesignal that is provided to the peaking-power-amplifier 814.

The splitter-phase-shifter 810 is implemented as a CLC circuit in thisexample, and includes a first-variable-capacitor 870, an inductor 872and a second-variable-capacitor 874. The first-variable-capacitor 870 isconnected between a phase-shifter-input-terminal and thereference-terminal 876. The inductor 872 is connected between thephase-shifter-input-terminal and a phase-shifter-output-terminal. Thesecond-variable-capacitor 874 is connected between thephase-shifter-output-terminal and the reference-terminal 876. The valuesof the first-variable-capacitor 870 and/or the second-variable-capacitor874 can be set/adjusted in order to provide a desired amount of phaseshift. In this way, the phase of the signal that is provided to thepeaking-power-amplifier 814 can be controlled.

In one example, the variable components (the variable resistors 866, 868and the variable capacitors 870, 874) can be implemented digitally byusing (n-type) MOSFETs as RF switches. For instance, an L-type DigitalStep Attenuator (DSA) can be used.

FIGS. 9a to 9c show various implementations of circuits that can be usedto provide the functionality of a variable component that is disclosedherein.

FIG. 9a shows an example of an L-type DSA (digital step attenuator) (forexample with NMOST devices). It will be appreciated that multiplesections of a series connection of a resistor and an NMOSFET switch canbe placed in parallel for both the shunt and series branch. The amountof parallelism may be chosen based on the amount of resolution, that is,the amount of bits that are considered appropriate for proper control.Additionally, the amount of parallelism can be designed based onpreferred coding, for example binary coding where the amount of parallelbranches equals the amount of bits, or thermometer coding where theamount of parallel branches equals 2 to the power of (amount of bits).An advantage of thermometer coding is that monotonicity is guaranteedwhich is beneficial for stability when the splitter-attenuator is partof a control loop.

FIG. 9b shows an example of a PI-type (π-type) DSAs can be used. ThePI-type DSA can include a shunt-R-series-R-shunt-R connection. In oneexample, the DSA can have 8 states (and therefore is 3 bit) and can havea step size of 0.5 dB.

FIG. 9c shows an example of a digitally adjustable phase shifter. Again,for example with NMOST devices. It will be appreciated that multiplesections of a series connection of a capacitor and an NMOSFET switch canbe placed in parallel.

Generally, various resistors, capacitors and switches can be used asunit devices having equal values, or can have different values.

Returning to FIG. 8, the splitter-phase-shifter 810 of FIG. 8 comprisesof a CLC topology with switchable parallel C branches. This topology canalso be implemented digitally. However, for applications that use largerbandwidths (400 MHz at 2 GHz), a CLC topology can have performancelimitations.

FIG. 10 shows an alternative implementation of a splitter-phase-shifter910, which is provided as a CLCCLC topology, where each CLC circuitshifts the phase by 45 degrees.

The splitter-phase-shifter 1010 includes a first-variable-capacitor1070, a first-inductor 1072, a second-variable-capacitor 1074, athird-variable-capacitor 1078, a second-inductor 1080 and afourth-variable-capacitor 1090. The first-variable-capacitor 1070 isconnected between a phase-shifter-input-terminal 1092 and areference-terminal 1076. The first-inductor 1072 is connected betweenthe phase-shifter-input-terminal 1092 and an intermediate-node 1096. Thesecond-variable-capacitor 1074 is connected between theintermediate-node 1096 and the reference-terminal 1076. Thethird-variable-capacitor 1078 is connected between the intermediate-node1096 and the reference-terminal 1076. The second-inductor 1080 isconnected between the intermediate-node 1096 and aphase-shifter-output-terminal 1094. The fourth-variable-capacitor 1090is connected between the phase-shifter-output-terminal 1094 and thereference-terminal 1076.

The splitter-phase-shifter 1010 can provide the following advantages. Itcan be less sensitive for process spread than the single-CLC topology.Also, it can be less frequency dependent and can lead to a moreorthogonal amplitude/phase control compared to the single-CLC topology.Also, good return losses can be achieved. For example, return lossesof >15 dB over a wider frequency range, i.e. bandwidth, can be provided.Alternatively, for a given bandwidth, the circuit can maintain betterreturn loss >15 dB over a wider phase control range. Therefore, thesplitter-phase-shifter 1010 can provide bandwidth and range benefits.

One or more of the examples disclosed herein can be considered as afully integrated Doherty amplifier, in which an input power splitter, amain amplifier, a peaking amplifier, and part of an output powercombiner are monolithically integrated. The other part of the outputpower combiner can be integrated on a laminate. The chip and laminatetogether reside within the same package. The input power splitter cancontain a digitally controlled phase shifter and attenuator. Bycontrolling the attenuator and phase shifter, the overall linearity ofthe Doherty amplifier can be optimized. The Doherty amplifier has anefficiency which is considerably larger than that of a class-ABamplifier, when amplifying signals which have a large peak-to-averagepower ratio like radio signals used in cellular communication systemsand wireless local area networks.

Examples disclosed herein can add value in terms of increasedintegration levels, including those examples that include a digital bus(e.g. SPI, I2C, MIPI) for example to implement adjustable-splittercontrol (amplitude and phase), bias-current control and system-readoutpossibilities (e.g. power level, temperature, over-voltage,under-voltage, over-current, over-temperature).

There can be advantages to providing the peaking-power-amplifier as aclass C amplifier, and making the class C threshold in the peaking pathprogrammable, since this determines the take-over point where thepeaking amplifier starts helping the main amplifier to deliver power tothe load. The class C threshold can be programmed based on the operatingfrequency, temperature, load impedance or supply voltage, for example.It can also be programmed based on the specific process spread that aproduct realization of the Doherty amplifier circuit has experienced.Hence, programming this peaking amplifier threshold can be part of thecalibration or self-calibration of the Doherty amplifier circuit.

One or more of the circuits disclosed herein can address the followingproblems: low-efficiency operation when amplifying signals having a highpeak-to-average-power-ratio, narrow-band signal transfer, nonlinearsignal amplification, low signal gain, and a bulky implementation.

When an ordinary single-chain class-AB amplifier would be used toamplify a signal having a high peak- to average-power ratio, then theaverage power level of the signal should be adjusted such that therewould be enough headroom to be able to amplify the large signal peakswithout introducing significant distortion. In other words: such asingle-chain amplifier should be operated in back-off mode. This canyield a low efficiency.

Examples disclosed herein contain two amplifier chains, instead of onlyone: which can be a main amplifier (class AB) and a peaking amplifier(class C). In this case, because the main amplifier is load-modulated bythe peaking amplifier, there may be no need to operate the mainamplifier in back-off mode since the signal peaks are handled by thepeaking amplifier. This can yield a high efficiency.

In examples disclosed herein, the signals generated by the main- andpeaking amplifiers are combined by an output power combiner whichintroduces impedance inversion in the main path and impedancetransformation in the peaking path. This impedance inversion andimpedance transformation can be designed in order to obtain a largesignal-transfer bandwidth.

In examples disclosed herein the input signal is split by means of anadjustable input power splitter into a first signal driving the mainamplifier and a second signal driving the peaking amplifier. This inputpower splitter introduces signal attenuation and phase shifting in thepeaking path. The amount of signal attenuation as well as phase shiftingcan be adjusted such as to optimize the overall linearity of the Dohertyamplifier.

In examples disclosed herein both the main- and peaking amplifier chainare equipped with multiple gain stages. This can yield a nearlysignal-strength independent loading on the input power splitter and itcan provide a large amount of power gain which is beneficial for thepower-added efficiency and can be convenient for the end user.

The instructions and/or flowchart steps in the above figures can beexecuted in any order, unless a specific order is explicitly stated.Also, those skilled in the art will recognize that while one example setof instructions/method has been discussed, the material in thisspecification can be combined in a variety of ways to yield otherexamples as well, and are to be understood within a context provided bythis detailed description.

In some example embodiments the set of instructions/method stepsdescribed above are implemented as functional and software instructionsembodied as a set of executable instructions which are effected on acomputer or machine which is programmed with and controlled by saidexecutable instructions. Such instructions are loaded for execution on aprocessor (such as one or more CPUs). The term processor includesmicroprocessors, microcontrollers, processor modules or subsystems(including one or more microprocessors or microcontrollers), or othercontrol or computing devices. A processor can refer to a singlecomponent or to plural components.

In other examples, the set of instructions/methods illustrated hereinand data and instructions associated therewith are stored in respectivestorage devices, which are implemented as one or more non-transientmachine or computer-readable or computer-usable storage media ormediums. Such computer-readable or computer usable storage medium ormedia is (are) considered to be part of an article (or article ofmanufacture). An article or article of manufacture can refer to anymanufactured single component or multiple components. The non-transientmachine or computer usable media or mediums as defined herein excludessignals, but such media or mediums may be capable of receiving andprocessing information from signals and/or other transient mediums.

Example embodiments of the material discussed in this specification canbe implemented in whole or in part through network, computer, or databased devices and/or services. These may include cloud, internet,intranet, mobile, desktop, processor, look-up table, microcontroller,consumer equipment, infrastructure, or other enabling devices andservices. As may be used herein and in the claims, the followingnon-exclusive definitions are provided.

In one example, one or more instructions or steps discussed herein areautomated. The terms automated or automatically (and like variationsthereof) mean controlled operation of an apparatus, system, and/orprocess using computers and/or mechanical/electrical devices without thenecessity of human intervention, observation, effort and/or decision.

It will be appreciated that any components said to be coupled may becoupled or connected either directly or indirectly. In the case ofindirect coupling, additional components may be located between the twocomponents that are said to be coupled.

In this specification, example embodiments have been presented in termsof a selected set of details. However, a person of ordinary skill in theart would understand that many other example embodiments may bepracticed which include a different selected set of these details. It isintended that the following claims cover all possible exampleembodiments.

1. A Doherty amplifier circuit comprising: a splitter having: asplitter-input-terminal for receiving an input signal; amain-splitter-output-terminal; and a peaking-splitter-output-terminal; amain-power-amplifier having a main-power-input-terminal and amain-power-output-terminal, wherein: the main-power-input-terminal isconnected to the main-splitter-output-terminal; and themain-power-output-terminal is configured to provide amain-power-amplifier-output-signal; a peaking-power-amplifier having apeaking-power-input-terminal and a peaking-power-output-terminal,wherein: the peaking-power-input-terminal is connected to thepeaking-splitter-output-terminal; and the peaking-power-output-terminalis configured to provide a peaking-power-amplifier-output-signal; anintegrated circuit; wherein the splitter, the main-power-amplifier andthe peaking-power-amplifier are provided by means of the integratedcircuit.
 2. The Doherty amplifier circuit of claim 1, wherein: thesplitter-input-terminal is connected to themain-splitter-output-terminal, and the splitter comprises asplitter-attenuator and a splitter-phase-shifter connected in cascadebetween the splitter-input-terminal and thepeaking-splitter-output-terminal.
 3. The Doherty amplifier circuit ofclaim 2, wherein the splitter-attenuator and the splitter-phase-shifterare provided on the integrated circuit.
 4. The Doherty amplifier circuitof claim 2, wherein the splitter-attenuator comprises a variablesplitter-attenuator that is configurable to apply a variable attenuationor amplification factor to the input signal.
 5. The Doherty amplifiercircuit of claim 4, wherein the variable splitter-attenuator comprises:a first-variable-resistor; and a second-variable-resistor wherein: thesecond-variable-resistor is connected between anattenuator-input-terminal and an attenuator-output-terminal; and thefirst-variable-resistor is connected between theattenuator-input-terminal and a reference-terminal.
 6. The Dohertyamplifier circuit of claim 2, wherein the splitter-phase-shiftercomprises a variable splitter-phase-shifter that is configurable toapply a variable phase shift to the input signal.
 7. The Dohertyamplifier circuit of claim 6, wherein the variablesplitter-phase-shifter comprises: a first-variable-capacitor, afirst-inductor, a second-variable-capacitor, a third-variable-capacitor,a second-inductor, and a fourth-variable-capacitor; wherein: thefirst-variable-capacitor is connected between aphase-shifter-input-terminal and a reference-terminal; thefirst-inductor is connected between the phase-shifter-input-terminal andan intermediate-node; the second-variable-capacitor is connected betweenthe intermediate-node and the reference-terminal; thethird-variable-capacitor is connected between the intermediate-node andthe reference-terminal; the second-inductor is connected between theintermediate-node and a phase-shifter-output-terminal; and thefourth-variable-capacitor is connected between thephase-shifter-output-terminal and the reference-terminal.
 8. The Dohertyamplifier circuit of claim 1, further comprising a controller that isconfigured to: receive one or more sensed-temperature-signals; and setcontrol parameters of the variable splitter-attenuator and/or thevariable splitter-phase-shifter based on the sensed-temperature-signals.9. The Doherty amplifier circuit of claim 8, wherein thesensed-temperature-signals are representative of: a temperature of theintegrated circuit, or a temperature of the main-power-amplifier or thepeaking-power-amplifier.
 10. The Doherty amplifier circuit of claim 1,further comprising: a transformer having a transformer-input-terminaland a transformer-output-terminal, wherein: thetransformer-input-terminal is connected to thepeaking-power-output-terminal; and the transformer-output-terminal isconfigured to provide a transformer-output-signal to a combining node.11. The Doherty amplifier circuit of claim 10, wherein the transformeris provided on the integrated circuit.
 12. The Doherty amplifier circuitof claim 1, further comprising: a package; and a laminate; wherein theintegrated circuit and the laminate are provided in the package.
 13. TheDoherty amplifier circuit of claim 12, further comprising amain-output-impedance-inverter connected between themain-power-output-terminal and a combining node, wherein themain-output-impedance-inverter comprises a CLC-inductor, afirst-CLC-capacitor and a second-CLC-capacitor, wherein: theCLC-inductor is connected between the main-power-output-terminal and thecombining node; the first-CLC-capacitor is connected between themain-power-output-terminal and a reference terminal; and thesecond-CLC-capacitor is connected between the combining node and thereference terminal.
 14. The Doherty amplifier circuit of claim 13,wherein the first-CLC-capacitor is provided on the integrated circuit,and wherein the CLC-inductor and the second-CLC-capacitor are providedon the laminate.
 15. The Doherty amplifier circuit of claim 1, whereinthe integrated circuit comprises a BiCMOS circuit.